1 Introduction

In order to replace the need of several antennas resonating at different frequencies, the small antenna must be equipped with on-demand frequency reconfigurability due to the rapid advancement of the various technologies [1]. The frequency ranges employed by common technologies including WLAN (5.15–5.85 GHz), WiMax (3.3–3.7 GHz), C-band (4–8 GHz), and X-band communication satellites (7.25–7.75 GHz) for space-to-earth communication and (7.9–8.4 GHz) for space-to-space communication [2]. Front ends for conventional RF communication systems include a receiver antenna, a mixing circuit, and a band-pass or bandstop filter. The enormous size of the circuit components, the bulky design, and the mismatch between the circuits components are some of the issues with these systems. Overcoming the disadvantage of employing distinct circuits is crucial for the progress of contemporary ultra-wideband communication schemes that are meant to be employed in contemporary communication networks. The so-called filtenna structure integrates the antenna with the necessary filters as a beautiful solution to this design dilemma [3].

High-speed ADCs are necessary for wideband antennas, but they use a lot of power and have larger quantization errors. Finally, because of constraints in the RF front-end components like amplifiers, oscillators, and mixers, the design for wideband operation compromises the transceiver performance. Narrowband frequency adjustable antennas, also known as tunable filter-antennas, are suitable candidates for use in order to sense the frequency spectrum due to all the aforementioned problems. This is due to the flexible frequency discrimination, wideband suppression of unwanted interference, gain flatness over the operating frequency band, less interference with the antenna's radiation characteristics, improved processing of down-converted signals, simplicity of implementation, and high performance of tunable band-pass filtering [4]. Due to their straightforward construction, patch and monopole antennas have been used by researchers for conformal applications. However, their disadvantageous size prevents them from being used for tiny devices [5, 6]. By incorporating dissimilar open-ended or short-ended stubs and slots into radiating constructions, the antenna size has been decreased. However, it adds structural complexity, which causes problems during fabrication [7]. In order to increase capacity, achieve diversity, improve the multipath effect, or minimize channel fading, MIMO techniques have been implemented in communication schemes [8,9,10,11]. Another way to increase capacity discrete wavelet transforms (DWT), DWTs are employed in orthogonal frequency division multiplexing (OFDM) systems to improve spectral efficiency. Therefore, wavelet transforms can be used to design antenna diversity scheme to improve performance of system and spectrum efficiency [12,13,14,15,16,17,18].

MIMO antenna design and deployment must take into account two key needs. These include constructing the antenna with the highest possible isolation between antenna elements and sizing it appropriately for portable applications. A good reception strategy is made possible by the strong isolation, which results in low mutual coupling (MC) between antennas [19]. It is well known that as the array size grows, the degree of coupling also grows. To lessen the MC in the MIMO antennas, many topologies and structures have been devised. The literature has discussed polarized orthogonal elements [20, 21], electromagnetic band-gap (EBG) structure [22], meander-line EBG [23], neutralization line [24, 25], stubs [26, 27], parasitic elements [28, 29], shorting pin [30], metamaterials with neutralization lines [31, 32], slots, or carbon black film [33]. Chen et al. [20] It describes a MIMO system with two orthogonal polarized antenna elements that has dimensions of 27 × 21 mm2 and a 22.5 dB isolation level. Authors of [22] introduced a quad-port double-side MIMO antenna. It is reliant on EBG structure and was created to enhance isolation. The MIMO system consists of four polygonal parts. Additionally, a \(30 \times 30\) mm2 FR4 substrate with a partly slotted EBG stub-ground is also comprised. A meander-line EBG construction has been employed in [23] to reduce MC between two \(32 \times 64\) mm2 antenna elements. So as to absorb the antenna interference, a carbon black layer was used in [33] to increase the amount of isolation. Achieved were \({S}_{21}< -15\mathrm{dB}\) and envelope correlation coefficients < 0.02. All prior designs, however, had problems with either excessive bulk or inadequate isolation.

The described works either have the drawbacks of solid substrates, greater dimensions, or set working frequencies, to wrap up this debate. As a result, this work makes two contributions. The first is the design and construction of a filtenna element with a circular base. This design is suggested because it creates a tiny antenna, conformal, and has on-demand frequency-switching between the thirteen different bands, which are used for C-band and X-band applications. It also gives a longer electrical line in a small space. For the purpose of preventing potential electromagnetic interference with other systems, the antenna provides harmonic suppression up to 12 GHz. The introduction of a dual-port MIMO filtenna with exceptional port-to-port separation follows. The MIMO construction is built using an orthogonal filtenna design to lessen element-to-element coupling. Inserting a metamaterial with a zigzag shape between adjacent parts further improves isolation. The MC effectively increases port-to-port isolation to more than 40 dB among the two ports across the majority of the working bandwidth when this zigzag shape is positioned between the antenna elements. This results in an overall size of \(80\times 45\) mm2. The structure of this essay is as follows. Method and experiment is described in Sect. 2. The tunable filtenna design process is described in Sect. 3. The MIMO filtenna system is discussed in Sect. 4 along with a demonstration of how to use a metamaterial to lower the isolation level. Section 5 presents the results and discussion. The paper is concluded in Sect. 6.

2 Method and experiment

Figure 1 shows the processes of the methodology for designing a dual-port MIMO filtenna. The MIMO filtenna and antenna element is simulated using the CST program. Each element in this MIMO filtenna is positioned orthogonally to the one next to it to improve isolation. To provide further isolation between the antennas ports, a DS between the antennas is added into the top layer.

Fig. 1
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Methodology steps to design the dual port MIMO filtenna

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3 Tunable filtenna design procedure

3.1 Wideband antenna design

A design for a WB circular patch antenna element based on partial ground structure is introduced in this subsection and is depicted in Fig. 2. A Rogers RT5880 substrate with a dielectric constant \({\varepsilon }_{r}\) of 2.2, a tan \(\delta\) of 0.0009, and a thickness \(h\) of 1.52 mm is used to print the antenna structure. The design was created using the software program microwave studio for computer simulation technology (CST). Table 1 contains the antenna's dimensions in millimeters. A feeding line with a width of \({W}_{\mathrm{f}}\) connects the patch to a 50 \(\Omega\) SMA connection. Figure 3a shows the simulation of the reflection coefficient \(|{S}_{11}|\) versus frequency. It is obvious that the suggested antenna covers a broad frequency range from 2.4 to more than 12 GHz for \(|{S}_{11}| \le -10\) dB.

Fig. 2
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Proposed WB antenna element structure a front view and b back view

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Table 1 Dimensions of the proposed WB antenna element in mm
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Fig. 3
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Simulated parameter of the WB antenna element. a \(|{S}_{11}|\), b efficiency, and c gain

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According to Fig. 3b, the simulated antenna efficiency spans the full operating frequency band and is between 80 and 99%. In addition, as presented in Fig. 3c, the antenna's simulated realized gain spans the working frequency band from 2.2 to 5.9 dB.

Figure 4 depicts the manufactured prototype of the suggested wideband antenna. As illustrated in Fig. 5, the suggested wideband antenna's scattering parameter \(|{S}_{11}|\) is measured using a Rohde & Schwarz ZVB 20 vector network analyzer. In the entire frequency range from 2.4 GHz to more than 12 GHz, the \(|{S}_{11}|\) is below the − 10 dB. The simulated and measured findings show a good degree of consistency. However, there is not much of a difference between the modeling and measurement results because of manufacturing tolerance levels.

Fig. 4
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Fabrication of the proposed wideband antenna a top view, b bottom view, and c S-Parameters measurement setup

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Fig. 5
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The simulated and measured reflection coefficient \(|{S}_{11}|\) of the proposed wide band antenna element

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3.2 Design of the proposed BPF-based DMS

In this subsection, the design of the planned BPF is presented. It is based on using of defected microstrip structure (DMS) technique. Firstly, a two tinny slit is etched at the midpoint of the stripe line, separating them from each other by 0.5mm which act as a fixed capacitance as displayed in Fig. 6. Secondly, a U-shaped slot of dimension 4.69 mm × 0.86 mm is etched at the middle part of strip line to control the coupling between the two transmission line sections. The purpose of the gaps is to allow the filter to have the “band-pass” feature by functioning as a parallel-series resonance.

Fig. 6
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Proposed DMS-BPF structure a front view and b back view

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The DMS construction can be expressed in terms of lumped elements by using circuit theory. The design is simulated with computer simulation technology (CST) software. So as to extract each of the different resonant frequencies, the S-parameters matrix obtained is then exported to the advanced design system (ADS). According to [34], the corresponding values for the capacitances and the inductances can be obtained from the next two equations.

$$C_{s,p} = \frac{{f_{{\text{c}}} }}{{200\pi \left( {f_{o}^{2} - f_{{\text{c}}}^{2} } \right)}},\;L_{s,p} = \frac{1}{{4\pi^{2} f_{o}^{2} C_{s,p} }}$$

where \({f}_{o}\) is the resonant frequency and \({f}_{\mathrm{c}}\) is the cutoff frequency of the designed filter.

The change in the values of \(C\) and, the filter operating frequency will change.

Figure 7 displays the equivalent circuit model for the DMS filter. The two gaps are modeled as parallel series LC and are represented by the inductor \({L}_{s}\) and the capacitor \({C}_{s}\), and the U-slot is modeled by \({L}_{p}\) and \({C}_{p}\). Figure 8a shows the simulated scattering parameter for DMS equivalent circuit in ADS simulation program. It is noticed that the behavior of the BPF is ideal and allows to pass frequency 7.6 GHz at specified value of of \(C\) and \(L\) as presented in Fig. 7. Figure 8b presents the simulated Scattering parameter for the proposed DMS BPF. It is noticed that S11 is below − 10dB at frequency 7.6 GHz and insertion loss is − 1.8dB.

Fig. 7
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a The equivalent circuit model for the DMS reconfigurable filter. b The Schematic circuit model in ADS simulation program

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Fig.8
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Simulated scattering parameter of a DMS equivalent circuit in ADS simulation program. b Proposed BPF-based DMS

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3.3 Frequency-reconfigurable filtering antenna

In this subsection, the design of the suggested tunable filtenna is presented. As shown in Fig. 9a, the BPF based on DMS is etched in the feed line of the WB antenna to create the filtenna. Impedance matching among the radiating patch and the BPF is made easier by this configuration while reducing system size and expense. To achieve reconfigurabitly and get variable frequencies, there are eight switches at different places in filtenna construction as shown in Fig. 9b. A comparison between the simulated \(\left|{S}_{11}\right|\) versus frequency of the antenna and the suggested filtenna is plotted in Fig. 10a. It is noted that the combination of the BPF with the antenna enables the wide band antenna to operate at a narrowband state with high selectivity. While the simulated \(\left|{S}_{11}\right|\) versus frequency of the tunable filtenna at various tuning switches is shown in Fig. 11. The achieved tuning frequency band is\(5.55\mathrm{ GHz}\), which spans the range from \(6.45\mathrm{ GHz}\) to more than\(12\mathrm{ GHz}\). The result of the changing state of switches, operational bandwidth (GHz), \(10\mathrm{ dB}\) BW, efficiency, and realized gain (dBi) of the suggested filtenna is depicted in Table 2. It is evident that the filter has high selectivity with narrow bandwidths ranging from \(17.3\mathrm{ to }94\mathrm{ MHz}\) over the entire tuning modes. Figure 10b, c provides clarification on the impact of the BPF's insertion loss, and switches, on the filtenna's efficiency and realized gain in\(dBi\). The realized gain and efficiency of the filtenna ranged from \(3\) to \(5.8\mathrm{ dBi}\) and 68–82.4%, respectively, within the tuning range.

Fig. 9
figure 9

a Front view of Suggested filtenna construction. b Front view of Suggested tunable filtenna construction with different switches

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Fig. 10
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Simulated parameter of the standalone antenna and antenna with filter. a \(|{S}_{11}|\), b efficiency, and c gain

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Fig. 11
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Simulated return losses \(|{S}_{11}|\) of the suggested filtenna at different states of the switches

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Table 2 The effectiveness of the suggested filtenna at different states of the switches
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A sample prototype of the planned filtenna at different states of the switches was fabricated using a standard chemical etching process to validate the design concept. Figure 12 shows the top side of the fabricated prototype at different states of the switches.

Fig. 12
figure 12

a Fabricated prototype of the proposed filtenna at different states of the switches. b The setup for measuring S-Parameters

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The comparison among simulated and measured reflection coefficients of the proposed antenna for various switching cases is plotted in Fig. 13.

Fig.13
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A comparison between the simulated and measured \(\left|{\mathrm{S}}_{11}\right|\) of filtenna at different states of the switches

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When switch S1 was kept on, while the other switches were kept in off-state (Case-00000001), the antenna resonated at 6.55 GHz, with a measured fractional bandwidth of 2.13%, ranging from 6.47 to 6.61GHz. On the other hand, when switches S4 and S6 were kept on, while the other switches were kept in off-state (Case-00101000), the antenna gave a resonance at 8.654 GHz, having measured |S11|<  − 10 dB impedance bandwidth of 4.15% (8.47–8.83GHz).Similarly, keeping switch S1 in on-state, when S7 was kept on, and while keeping the others in off-state, then the antenna operated at 9.837GHz for Case-01000001. Commonly, an acceptable agreement between the measured and simulated results was detected for all switching states. However, the slight discrepancy among predicted and measured results is due to fabrication and measurement setup tolerances. Table 3 compares the performance of the proposed filtenna to a recently published study. It is evident that the suggested filtenna has a more compact construction and a wider tunable band when compared to existing filtennas. Mechanically switches is used, which is another advantage.

Table 3 Comparison between the suggested filtenna and published studies
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4 Design of the MIMO filtenna

In a communication system implementing diversity, the same data are sent over independent fading paths. The signals received over the independent paths are then combined in such a way that the fading of the resultant signal is reduced. A wireless communication system equipped with diversity antennas leads to improved capacity and quality of the wireless channel.

In this section, a pattern diversity antenna with a frequency tuning feature (MIMO filtenna) is presented. Figure 14a shows the MIMO filtenna's structural layout. Two filtenna elements are used in the MIMO system, and they are positioned orthogonally with respect to one another. To perform well, this organization lowers the level of isolation. In the design, port 1 and port 2 are the MIMO ports. Consequently, MCs between port i and port j (Sij), i, j = 1, 2, i ≠ j are minimized. Figure 14b shows the scattering parameter of the proposed MIMO filtenna without DS. S11 is − 45 db at resonance frequency 7.55 GHz and S12 is − 30 dB.

Fig.14
figure 14

Geometry of the proposed MIMO filtenna without DS. a topview. b bottomview. c S-Parameters of the proposed MIMO filtenna without DS

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5 Results and discussion

Figure 15 depicts the MIMO structure's surface current distribution when one port is activated and the other is closed. One may observe that when one port is activated, the other ports also experience a certain amount of current. As a result, coupling requires greater reduction, which will enhance isolation. To achieve this, a special decoupling structure (DS) to support isolation between the antenna parts is recommended.

Fig.15
figure 15

The suggested MIMO elements' surface current distribution when a excited port 1 and b excited port 2

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The core of the DS idea is the creation of a metamaterial structure (like zigzag) that is sandwiched between the MIMO antenna parts, supposedly suppressing all energy. To reduce the implantation area and achieve good isolation between the MIMO parts, the high impedance wire is meandering in a zigzag pattern. Figure 16a presents the geometry of the final structure of the proposed two-port MIMO filtenna together with specific parameters. The two-port MIMO filter described is \(80 \times 45\) mm2 in size overall. Figure 16b shows the scattering parameter of the proposed MIMO filtenna without DS. S11 is − 45 db at resonance frequency 7.55 GHz and S12 is − 65 dB.

Fig.16
figure 16

a The proposed MIMO filtenna's geometry with the DS. \({\mathrm{W}}_{\mathrm{MIMO}}=80\mathrm{mm}\), \({\mathrm{L}}_{\mathrm{MIMO}}=45\mathrm{mm}\), \({\mathrm{W}}_{\mathrm{z}}={\mathrm{L}}_{\mathrm{z}}=2.5\mathrm{mm}\). b The suggested MIMO filtenna's S-parameters with DS

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A comparison of isolation levels for instances with and without the DS is shown in Fig. 17. The proposed DS significantly minimizes the MC between antenna ports, which is noteworthy. Only port 1's findings are shown due of the symmetry. At 7.55 GHz, the coupling between port 1 and port 2 (S12) is less than − 20 dB for the case without DS. At 7.55 GHz, an improvement of around 40 dB is made with the DS, as depicted in Fig. 17. The distribution of surface current is examined for more proof. As seen in Fig. 18, Port (1) is individually energized at 7.55 GHz to study the DS effect. It has been shown that adding the proposed DS in between the antenna elements greatly reduces the current drift between them.

Fig. 17
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Comparison of mutual coupling with and without DS

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Fig. 18
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Surface current distribution at 7.55 GHz for the proposed MIMO filtenna when a Port (1) is excited, b Port (2) is excited

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5.1 MIMO performance study

5.1.1 ECC

In a MIMO system, the relationship between antenna elements is reflected by the ECC [36, 41]. For the estimation of the ECC, the \({S}_{ij}\) and \({S}_{ji}\) are used. For optimal MIMO system performance, the ECC should be less than 0.5 [37, 39]. The following is an estimation of the ECC [40]:

$$\mathrm{ECC}=\frac{{\left|{S}_{11}^{*}{S}_{12}+{S}_{21}^{*}{S}_{22}\right|}^{2}}{\left(1-{\left|{S}_{11}\right|}^{2}+{\left|{S}_{21}\right|}^{2}\right)\left(1-{\left|{S}_{22}\right|}^{2}+{\left|{S}_{12}\right|}^{2}\right)}$$

The far-field radiation pattern can also be used to determine the ECC in the manner shown below. [40]:

$$ECC=\frac{{\left|\int \int \left[{\overrightarrow{F}}_{1}\left(\theta ,\phi \right)*{\overrightarrow{F}}_{2}\left(\theta ,\phi \right)\right]\mathrm{d\Omega }\right|}^{2}}{\iint {\left|\left[{\overrightarrow{F}}_{1}\left(\theta ,\phi \right)\right]\right|}^{2}\mathrm{d\Omega }\iint {\left|\left[{\overrightarrow{F}}_{2}\left(\theta ,\phi \right)\right]\right|}^{2}\mathrm{d\Omega }}$$

where \({\overrightarrow{F}}_{1}\left(\theta ,\phi \right)\) and \({\overrightarrow{F}}_{2}\left(\theta ,\phi \right)\) are the field patterns of two filtenna elements, when port (1) and port (2) are activated, respectively. \(\theta ,\phi\) and \(\Omega\) are solid angels, and the Hermitian product is indicated by the asterisk (*). The simulated ECC of the suggested MIMO system for the two instances with and without the DS is shown in Fig. 19a. Without the DS, the MIMO antenna's maximum ECC is equal to 0.00012, whereas with the DS, it is equivalent to 2.4e−6.

Fig. 19
figure 19

The simulated MIMO parameters a ECC, b DG, and c TARC

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5.1.2 Diversity gain (DG)

The DG is alternative pointer for the diversity level. A high value is necessary for effective isolation. (5) [42] can be used to estimate it.

$$\mathrm{DG}=\sqrt{1-{\mathrm{ECC}}^{2}}$$

The DG for the proposed MIMO filtenna is greater than 9.9 in both circumstances, as shown in Fig. 19b.

5.1.3 Total active reflection coefficient (TARC)

The square root of the entire reflected power divided by the square root of the entire incident power is known as the TARC. Equation (6) is used to estimate it [24], and [43]. S-parameter curves converge as illustrated in Fig. 19c by inferring to the first port signal phase and altering the remaining port signal phases by three distinct values: 90°, 180°, and 270°. Over the whole working bandwidth, various signal phase variations perform effectively.

$${\Gamma }_{a}^{t}=\frac{\sqrt{{\left|({S}_{11}+{S}_{12}{e}^{j\theta })\right|}^{2}+{\left|({S}_{21}+{S}_{22}{e}^{j\theta })\right|}^{2}}}{2}$$

5.2 Manufacture and measurement

Fabricated and measured MIMO filtenna proposed. The fabricated MIMO filtenna is depicted in Fig. 20a. The measuring setup for the measurements of the S-parameters and the radiation pattern is shown in Fig. 20b, c. A comparison of the simulated and measured VSWR and scattering characteristics is shown in Fig. 21. The simulation and measurements show excellent agreement. Figure 22 depicts the radiation patterns for the x–y and y–z planes at 7.55GHz, respectively. In the y–z plane, the radiation pattern is infinity shaped, whereas in the x–y plane, it is omnidirectional. Figure 23 presents an illustration of gain over frequency. A gain of 6 dBi maximum is attained.

Fig. 20
figure 20

a A MIMO antenna that was made. b The setup for measuring S-Parameters. c The setup for measuring radiation patterns

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Fig. 21
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An evaluation of the simulated and measured a VSWR, b Scattering parameters

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Fig. 22
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The proposed MIMO filtenna's simulated and measured y–z and x–y radiation patterns a y–z at 7.55 GHz, b x–y at 7.55 GHz

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Fig. 23
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Simulated and measured gain of proposed MIMO filtenna

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Table 4 provides a comparison of previously published literary works. The best element-to-element isolation, highest peak gain, and comparatively small size are displayed by the proposed MIMO filtenna. Also, among all literary works, the ECC and DG are the best.

Table 4 Comparison among the suggested MIMO filtenna and the literary work
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6 Conclusion

A brand-new, small, circular antenna with an impedance bandwidth of 2.4 to more than 12 GHz has been created. For C-band applications, a filtenna with tunable frequencies from 6.45 to 8 GHz and from 8 to 12 GHz for X-band applications was built. A dual-port MIMO antenna with a surface area of 80 × 45 mm2 has been used. The isolation between ports has been improved using orthogonal polarization and a metamaterial. It has proven possible to achieve a MIMO filtenna with an impedance bandwidth of 7.4–7.75 GHz, an isolation level of 76 dB, a peak gain of 6 dBi, and an ECC lower than 2.4e–6. The MIMO filtenna has been made up and measured. The modeling and measurements have achieved good agreement.