The MicroCap12 System of Circuit Modeling

Abstract

The effectiveness of various methods for analyzing individual nodes of a RF front end based on the use of linear and nonlinear AE models is discussed and compared, limitations are indicated, and current trends in the design of radio engineering nodes and devices are considered. The expediency of using digital methods of analysis is confirmed while maintaining the physical description of the ongoing processes and the AE models used in the MicroCap environment, as well as the usual forms of presentation of the results: process diagrams at various points of the circuit in the time domain, spectral, and amplitude-frequency characteristics.

The high accuracy and efficiency of the results of the numerical analysis of devices based on the use of nonlinear AE models is shown, but only under the condition that the parameters of the Spice AE models coincide with the data published in the reference books for the corresponding components. Of course, in this case, the condition that the applied descriptions of the AE models fit into the admissible limits of its spread, given in reference books of average physical parameters, with real samples, must be met.

Such an analysis makes it possible to obtain information that is not attainable when using linear models of the AE, when its operating mode for direct current changes or the AE switches from a linear mode in terms of the input action level to a nonlinear one, for example, when assessing the harmonic factor and the level of intermodulation distortion.

Appendix

6.1.1 Application 6.1 Calculation of small-signal Y-parameters of an active four-port network using the MicroCap program

The task of the designers of the radio receiving equipment (amplifiers, frequency converters), regardless of the degree of integration and the operating frequency range, is to achieve the required indicators, the most important of which are a given operating frequency band, gain, permissible nonuniformity of frequency response in the operating band, and nonlinear distortion coefficient.

Thus, the task of designing, for example, a resistor cascade on a BT, includes two stages: ensuring the operation of the AE in direct current and the main characteristics of the cascade.

• 1. Checking the AE mode for direct current.

The low level of input signals allows the use of A mode for AE. The choice of the BT mode in receivers is mainly determined by the place where the cascade is switched on in the RF front end and, for example, for its first stages it should provide Ic0 = 2 mA at Uce = 2.5 V.

• 2. Calculation of small-signal parameters of BT to estimate the gain and other parameters of the stage.

If in the first case it is necessary to use a real circuit of stage, then to calculate the gain, it is necessary to know the small-signal parameters of the BT for the selected transistor manner to turn on. In the equivalent circuit of the two-port by the system of Y-parameters, coefficients of the equation system, defining its own and mutual conductance, are calculated at short circuit of input and output terminals in AC .

As an example, we will calculate the input conductivity of a BT small signal, switched on according to the scheme with OE, using the electrical circuit in the figure Appendix 6.1 in the MicroCap environment.

The “short circuit” mode for alternating current at the BT output is ensured by switching on a large capacitor C8. Internal resistance in power supply models (V11 and V10) is selected to be zero (ideal EMF sources). Inductors L5 and L6 exclude the influence of external circuits connected to the transistor on its performance, as well as the signal source (V12). BT properties are described by its PSPICE model:

• .MODEL 2T325V NPN (BF = 321.5 BR = 1.78 CJC = 2.958p CJE = 3.42p IKF = 87.77 m IKR = 606.8 m IS = 9.164f ISC = 100f ISE = 87.74f ITF = 300 m NC = 1.744 NE=1.473 RB=31 RC=0.2997 TF=122.2p TR=8.891n VAF=87 VAR=45 VTF=25 XTB=1.5 XTF=2).

After numbering the nodes and executing the commands Analysis → Dynamic DC Limits, select the V icon from the drop-down sub-menu, and display the voltage values in the circuit nodes on the monitor screen (Fig. 6.50).

Fig. 6.50

Auxiliary circuit for calculating small-signal Y-parameters of BT (input conductivity, capacitance and direct transmission conductivity)

If they differ from the analysis conditions, it is necessary to correct the source voltage V11.

For a fixed value of Uce = 2.5 V, the current at the operating point is set by varying the voltage of the bias source (V10) connected to the BT base. For a source voltage of V10 = 0.676 mV, the required collector current is provided: Ic0 = 2 mA (Fig. 6.51).

Fig. 6.51

Distribution of direct current (DC) in the measuring circuit

To perform calculations of the input and conductivity of the direct transmission, it is necessary to execute in the circuit window: Analysis → AC… → Limits, go to the menu for setting the conditions of the circuit analysis: AC Analysis Limits (Fig. 6.52).

Fig. 6.52

Analysis conditions, methods, and limits of presentation of obtained results

As follows from the description of Dynamic AC limits submenu (Fig. 6.44), the output parameters of the simulation are the module and phase of voltages and current of the circuit components. This allows you to find the actual part of the conductivity of the corresponding ports of the circuit. Considering that the reactive component of the input (and output) conductivity of BT has a capacitive character, therefore, in the line (Fig. 6.53b) there is C11, its input capacitance. In the selected example, in the frequency range from 10 kHz to 1 GHz, the real part of the input conductivity Re (Y11) BT (denoted as g11 in Fig. 6.53a), the input capacitance of the transistor (Fig. 6.53b), the input conductivity module │Y11│ (Fig. 6.53c), and the direct transmission conductivity module │Y21│ (Fig. 6.53d) are calculated.

Fig. 6.53

Calculation of the components of the Y-parameter matrix of the BT. (a) the real part of the input conductivity, (b) the input capacitance, (c) the modulus of the input conductivity, (d) the modulus of the direct transmission

Calculation in the MicroCap environment of the input conductivity and input capacitance of the DC allows you to obtain values that are often used when manually calculating the parameters of the amplifying stages for the selected transistor DC mode and a specific operating frequency.

The value of any indices on the operation frequency is defined with the help of the cursor by extracting the interested frequency region in the sub-menu AC Analysis or by reduction of the region of analyzed parameters in columns Limits: XRange and YRange.

6.1.2 Application 6.2 Calculate the mean-square value of the noise voltage in input and output of the resistor stage (Fig. 6.54)

The electrical diagram of the stage includes the bipolar transistor of NPN type, parameters of which nonlinear PSPICE model includes:

• .MODEL KT316D NPN (BF = 136.5 BR = 0.6577 CJC = 4.089p CJE = 1.16p IKF = 97.23 m IKR = 120 m EG = 1.11 FC=0.5 IS = 2.753f ISC = 15.5p ISE = 12.8p ITF = 151 m NE =2.496 RB = 70.6 RBM = 70.6 RC = 8.35 TF = 78.97p NC = 2 MJC = 0.33 MJE = 0.33 TR = 27.84n VAF = 96 VAR = 55 VJC = 650 m VJE = 690 m VTF = 25 XTB = 1.5 XTI = 3 XTF = 2).

Fig. 6.54

Schematic diagram of a resistor stage on BT

The mode of operation of the KT316D transistor, switched on according to the scheme with common emitter (I_c0 = 2 mA, U_ce0 = 2.7 V), must be provided by the previously calculated or selected value of components. At the first stage of modeling, the correspondence of the real position of the working point of the BT and the one obtained in the preliminary calculation is checked. To do this, in the schematics window, the following is performed sequentially: Analysis > Dynamic DC…

and on the Dynamic DC Limits submenu, icons are selected that display the currents in circuits and voltages in nodes. By clicking OK, we get the values of the collector current and the voltage calculated by the program. If the difference between the selected working point and the one obtained during modeling is more than 20%, the correction of the circuit components is performed. The total source V2 provides the power supply for collector circuit and with the help of а divider on R11 and R12 the bias voltage on the base of the bipolar transistor. The resistor R13 is the component of the negative feedback circuit in DC , which improves the stability of operation mode of the bipolar transistor and its gain.

In the circuit input, the generator of the harmonic oscillation V1 (GIN) connects, which is described by the model. MODEL GIN SIN (A = 0.01 F = 5MEG).

Noise indices of the amplifying stage take into account: thermal noises, flicker and shot noises. The circuit resistors, which play the auxiliary role, and the spurious resistors of the bipolar transistor are the sources of thermal noises.

Flickering noises are caused by processes occurring in bipolar transistors caused by defects in the crystal lattice in the semiconductor structure, causing random generation and recombination of charge carriers, the presence of traps in the base structure and

surface states. A distinctive feature of such noises is their spectral characteristic of the form ~1/fⁿ, where n = 2, when their intensity increases significantly near zero frequency. The noise of the shot usually reflects random changes in the injection and recombination current in semiconductors. The model of the source of such noise caused by the uneven flow of charge carriers through the collector-base junction is a current generator controlled by the base current, which, in turn, is determined by the degradation of the semiconductor structure.

The noise analysis is based on the measurement of the total contribution of all sources in the output of the circuit for the node (6), indicated in the line Noise output of the sub-menu AC Analysis Limits (Fig. 6.55). The spectral density of the mean-square value of the noise voltage (RMS, root mean square, dimension V/$\sqrt{Hz}$) is defined through the noise power spectral density in this node (V²/Hz).

Fig. 6.55

Calculated noise characteristics at the amplifier input (solid line) and the total noise at its output (marked curve)

This plot presents the spectral density of the mean-square EMF in dB in the input (inoise, solid curve) and in output (onoise) of the circuit marked by points.

The input noise is calculated through the power value of the output noise by division on the frequency-dependent transfer function from the input (mentioned in the line Noise input, V1) to the output.

A feature of the calculation of noise indicators in the frequency domain is the use of a low-signal model of the circuit, which does not allow to simultaneously perform traditional analysis by the method of variable states when a harmonic signal of a much higher level is used as source.